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1 MHz to 10 GHz, 40 dB Log Detector/Controller AD8319
FEATURES
Wide bandwidth: 1 MHz to 10 GHz High accuracy: 1.0 dB over temperature >40 dB dynamic range up to 8 GHz Stability over temperature 0.5 dB Low noise measurement/controller output VOUT Pulse response time: 8/10 ns (fall/rise) Small footprint 2 mm x 3 mm CSP package Supply operation: 3.0V to 5.5V @ 22 mA Fabricated using high speed SiGe process
FUNCTIONAL BLOCK DIAGRAM
VPOS TADJ
GAIN BIAS
SLOPE
I
V
VSET
I DET INHI INLO DET DET DET
V
VOUT
CLPF
COMM
APPLICATIONS
RF transmitter PA setpoint control and level monitoring Power monitoring in radiolink transmitters RSSI measurement in base stations, WLAN, WiMAX, radar
Figure 1.
GENERAL DESCRIPTION
The AD8319 is a demodulating logarithmic amplifier, capable of accurately converting an RF input signal to a corresponding decibel-scaled output. It employs the progressive compression technique over a cascaded amplifier chain, each stage of which is equipped with a detector cell. The device can be used in either measurement or controller modes. The AD8319 maintains accurate log conformance for signals of 1 MHz to 8 GHz and provides useful operation to 10 GHz. The input dynamic range is typically 40 dB (re: 50 ) with error less than 1 dB. The AD8319 has 8/10 ns response time (fall time/rise time) that enables RF burst detection to a pulse rate of beyond 50 MHz. The device provides unprecedented logarithmic intercept stability vs. ambient temperature conditions. A supply of 3.0 V to 5.5 V is required to power the device. Current consumption is typically 22 mA, and it decreases to 200 A when the device is disabled. The AD8319 can be configured to provide a control voltage to a power amplifier or a measurement output from the VOUT pin. Because the output can be used for controller applications, special attention has been paid to minimize wideband noise. In this mode, the setpoint control voltage is applied to the VSET pin. The feedback loop through an RF amplifier is closed via VOUT, the output of which regulates the amplifier's output to a magnitude corresponding to VSET. The AD8319 provides 0 V to (VPOS - 0.1 V) output capability at the VOUT pin, suitable for controller applications. As a measurement device, VOUT is externally connected to VSET to produce an output voltage VOUT that is a decreasing linear-in-dB function of the RF input signal amplitude. The logarithmic slope is -22 mV/dB, determined by the VSET interface. The intercept is +15 dBm (re: 50 , CW input) using the INHI input. These parameters are very stable against supply and temperature variations. The AD8319 is fabricated on a SiGe bipolar IC process and is available in a 2 mm x 3 mm, 8-lead LFCSP_VD package for an operating temperature range of -40oC to +85oC.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c)2005 Analog Devices, Inc. All rights reserved.
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AD8319 TABLE OF CONTENTS
Features .............................................................................................. 1 Applications....................................................................................... 1 General Description ......................................................................... 1 Functional Block Diagram .............................................................. 1 Revision History ............................................................................... 2 Specifications..................................................................................... 3 Absolute Maximum Ratings............................................................ 5 ESD Caution.................................................................................. 5 Pin Configuration and Function Descriptions............................. 6 Typical Performance Characteristics ............................................. 7 Theory of Operation ...................................................................... 10 Using the AD8319 .......................................................................... 11 Basic Connections ...................................................................... 11 Input Signal Coupling................................................................ 11 Output Interface ......................................................................... 11 Setpoint Interface ....................................................................... 11 Temperature Compensation of Output Voltage..................... 12 Measurement Mode ................................................................... 12 Setting the Output Slope in Measurement Mode .................. 13 Controller Mode......................................................................... 13 Output Filtering.......................................................................... 15 Operation Beyond 8 GHz ......................................................... 15 Evaluation Board ............................................................................ 16 Outline Dimensions ....................................................................... 18 Ordering Guide .......................................................................... 18
REVISION HISTORY
10/05--Revision 0: Initial Version
Rev. 0 | Page 2 of 20
AD8319 SPECIFICATIONS
VPOS = 3 V, CLPF = 1000 pF, TA = 25C, 52.3 termination resistor at INHI, unless otherwise noted. Table 1.
Parameter SIGNAL INPUT INTERFACE Specified Frequency Range DC Common-Mode Voltage MEASUREMENT MODE f = 900 MHz Input Impedance 1 dB Dynamic Range Maximum Input Level Minimum Input Level Slope 1 Intercept1 Output Voltage: High Power In Output Voltage: Low Power In f = 1.9 GHz Input Impedance 1 dB Dynamic Range Maximum Input Level Minimum Input Level Slope1 Intercept1 Output Voltage: High Power In Output Voltage: Low Power In f = 2.2 GHz Input Impedance 1 dB Dynamic Range Maximum Input Level Minimum Input Level Slope1 Intercept1 Output Voltage: High Power In Output Voltage: Low Power In f = 3.6 GHz Input Impedance 1 dB Dynamic Range Maximum Input Level Minimum Input Level Slope1 Intercept1 Output Voltage: High Power In Output Voltage: Low Power In Conditions INHI (Pin 1) Min 0.001 VPOS - 0.6 VOUT (Pin 5) shorted to VSET (Pin 4), sinusoidal input signal RTADJ = 18 k TA = +25C -40C < TA < +85C 1 dB error 1 dB error -25 12 PIN = -10 dBm PIN = -40 dBm RTADJ = 8 k TA = +25C -40C < TA < +85C 1 dB error 1 dB error -25 10 PIN = -10 dBm PIN = -35 dBm RTADJ = 8 k TA = +25C -40C < TA < +85C 1 dB error 1 dB error 1500||0.33 40 40 -3 -43 -22 15 0.57 1.25 950||0.38 40 40 -4 -44 -22 13 0.53 1.19 810||0.39 40 40 -5 -45 -22 13 0.5 1.18 300||0.33 40 36 -6 -46 -22 10 0.46 1.14 ||pF dB dB dBm dBm mV/dB dBm V V ||pF dB dB dBm dBm mV/dB dBm V V ||pF dB dB dBm dBm mV/dB dBm V V ||pF dB dB dBm dBm mV/dB dBm V V Typ Max 10 Unit GHz V
-19.5 21
-19.5 20
PIN = -10 dBm PIN = -35 dBm RTADJ = 8 k TA = +25C -40C < TA < +85C 1 dB error 1 dB error
PIN = -10 dBm PIN = -40 dBm
Rev. 0 | Page 3 of 20
AD8319
Parameter f = 5.8 GHz Input Impedance 1 dB Dynamic Range Maximum Input Level Minimum Input Level Slope1 Intercept1 Output Voltage: High Power In Output Voltage: Low Power In f = 8.0 GHz Input Impedance 1 dB Dynamic Range Maximum Input Level Minimum Input Level Slope 2 Intercept2 Output Voltage: High Power In Output Voltage: Low Power In OUTPUT INTERFACE Voltage Swing Output Current Drive Small Signal Bandwidth Output Noise Fall Time Fall Time Rise Time Rise Time Video Bandwidth (or Envelope Bandwidth) VSET INTERFACE Nominal Input Range Logarithmic Scale Factor Input Resistance TADJ INTERFACE Input Resistance Disable Threshold Voltage POWER INTERFACE Supply Voltage Quiescent Current vs. Temperature Disable Current
1 2
Conditions RTADJ = 500 TA = +25C -40C < TA < +85C 1 dB error 1 dB error
Min
Typ 110||0.05 40 40 -3 -43 -22 15 0.57 1.25 28||0.79 40 31 -1 -41 -22 20 0.67 1.34 VPOS - 0.1 10 10 140 90 18 6 20 10 50
Max
Unit ||pF dB dB dBm dBm mV/dB dBm V V ||pF dB dB dBm dBm mV/dB dBm V V V mV mA MHz nV/Hz ns ns ns ns MHz
PIN = -10 dBm PIN = -40 dBm RTADJ = open TA = +25C -40C < TA < +85C 1 dB error 1 dB error
PIN = -10 dBm PIN = -40 dBm VOUT (Pin 5) VSET = 0 V; RFIN = open VSET = 1.5 V; RFIN = open VSET = 0 V; RFIN = open RFIN = -10 dBm; from CLPF to VOUT RF Input = 2.2 GHz, -10 dBm, fNOISE = 100 kHz, CLPF = open Input level = no signal to -10 dBm, 90% to 10%; CLPF = 8 pF Input level = no signal to -10 dBm, 90% to 10%; CLPF = open; ROUT = 150 Input level = -10 dBm to no signal, 10% to 90%; CLPF = 8 pF Input level = -10 dBm to no signal, 10% to 90%; CLPF = open; ROUT = 150
VSET (Pin 4) RFIN = 0 dBm; measurement mode RFIN = -40 dBm; measurement mode RFIN = -20 dBm; controller mode; VSET = 1 V TADJ (Pin 6) TADJ = 0.9 V, sourcing 50 A TADJ = Open VPOS (Pin 7) 3.0 18 -40C TA +85C TADJ = VPOS
0.35 1.23 -45 40 40 VPOS - 0.4 5.5 30
V dB/V k k
22 60 200
V mA A/C A
Slope and intercept are determined by calculating the best-fit line between the power levels of -40 dBm and -10 dBm at the specified input frequency. Slope and intercept are determined by calculating the best-fit line between the power levels of -34 dBm and -16 dBm at 8.0 GHz.
Rev. 0 | Page 4 of 20
AD8319 ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Supply Voltage: VPOS VSET Voltage Input Power (Single-Ended, Re: 50 ) Internal Power Dissipation JA Maximum Junction Temperature Operating Temperature Range Storage Temperature Range Lead Temperature (Soldering 60 sec) Rating 5.7 V 0 to VPOS 12 dBm 0.73 55C/W 125C -40C to +85C -65C to +150C 260C
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
Rev. 0 | Page 5 of 20
AD8319 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
INHI 1 COMM 2 CLPF 3 VSET 4 8 INLO
AD8319
TOP VIEW (Not to Scale)
7 VPOS 6 TADJ 5 VOUT
05705-002
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No. 1 2 3 4 5 Mnemonic INHI COMM CLPF VSET VOUT Description RF Input. Nominal input range of -50 dBm to 0 dBm, re: 50 ; ac-coupled RF input. Device Common. Connect to a low impedance ground plane. Loop Filter Capacitor. In measurement mode, this capacitor sets the pulse response time and video bandwidth. In controller mode, the capacitance on this node sets the response time of the error amplifier/integrator. Setpoint Control Input for Controller Mode or Feedback Input for Measurement Mode. Measurement and Controller Output. In measurement mode, VOUT provides a decreasing linear-in dB representation of the RF input signal amplitude. In controller mode, VOUT is used to control the gain of a VGA or VVA with a positive gain sense (increasing voltage increases gain). Temperature Compensation Adjustment. Frequency-dependent temperature compensation is set by connecting a ground-referenced resistor to this pin. Positive Supply Voltage: 3.0 V to 5.5 V. RF Common for INHI. AC-coupled RF common. Internally connected to COMM; solder to a low impedance ground plane.
6 7 8
TADJ VPOS INLO Paddle
Rev. 0 | Page 6 of 20
AD8319 TYPICAL PERFORMANCE CHARACTERISTICS
VPOS = 3 V; T = 25C, -40C, +85C; CLPF = 1000 pF; unless otherwise noted. Colors: 25C Black; -40C Blue; 85C Red Error is calculated by using the best-fit line between PIN = -40 dBm and PIN = -10 dBm at the specified input frequency, unless otherwise noted.
2.00 1.75 1.50 1.25
VOUT (V)
2.0 1.5 1.0
ERROR (dB)
2.00 1.75 1.50 1.25
VOUT (V)
2.0 1.5 1.0
ERROR (dB)
05705-008
0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
1.00 0.75 0.50 0.25 0 -60
1.00 0.75 0.50 0.25 0 -60
05705-003
Figure 3. VOUT and Log Conformance vs. Input Amplitude at 900 MHz, RTADJ = 18 k
2.00 1.75 1.50 1.25
VOUT (V)
Figure 6. VOUT and Log Conformance vs. Input Amplitude at 3.6 GHz, RTADJ = 8 k
2.00 1.75 1.50
ERROR (dB)
2.0 1.5 1.0 0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
2.0 1.5 1.0
ERROR (dB) ERROR (dB)
05705-007
1.25
VOUT (V)
0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
1.00 0.75 0.50 0.25 0 -60
1.00 0.75 0.50 0.25 0 -60
Figure 4. VOUT and Log Conformance vs. Input Amplitude at 1.9 GHz, RTADJ = 8 k
2.00 1.75 1.50 1.25
VOUT (V)
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Figure 7. VOUT and Log Conformance vs. Input Amplitude at 5.8 GHz, RTADJ = 500
2.00 1.75 1.50
ERROR (dB)
2.0 1.5 1.0 0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
2.0 1.5 1.0 0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
1.25
VOUT (V)
1.00 0.75 0.50 0.25 0 -60
1.00 0.75 0.50 0.25 0 -60
Figure 5. VOUT and Log Conformance vs. Input Amplitude at 2.2 GHz, RTADJ = 8 k
Rev. 0 | Page 7 of 20
05705-005
Figure 8. VOUT and Log Conformance vs. Input Amplitude at 8.0 GHz, RTADJ = Open, Error Calculated from PIN = -34 dBm to PIN = -16 dBm
05705-006
AD8319
2.00 1.75 1.50 1.25
VOUT (V)
2.00 1.75 1.50
ERROR (dB)
2.00 1.75 1.50 1.25
VOUT (V)
2.0 1.5 1.0
ERROR (dB)
05705-014
1.25 1.00 0.75 0.50 0.25 0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
1.00 0.75 0.50 0.25 0 -60
1.00 0.75 0.50 0.25 0 -60
Figure 9. VOUT and Log Conformance vs. Input Amplitude at 900 MHz, Multiple Devices, RTADJ = 18 k
2.00 1.75 1.50 1.25
VOUT (V)
05705-009
Figure 12. VOUT and Log Conformance vs. Input Amplitude at 3.6 GHz, Multiple Devices, RTADJ = 8 k
2.00 1.75 1.50
ERROR (dB)
2.0 1.5 1.0 0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
2.0 1.5 1.0
ERROR (dB) ERROR (dB)
05705-013
1.25
VOUT (V)
0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
1.00 0.75 0.50 0.25 0 -60
1.00 0.75 0.50 0.25
Figure 10. VOUT and Log Conformance vs. Input Amplitude at 1.9 GHz, Multiple Devices, RTADJ = 8 k
2.00 1.75 1.50 1.25
VOUT (V)
05705-010
0 -60
Figure 13. VOUT and Log Conformance vs. Input Amplitude at 5.8 GHz, Multiple Devices, RTADJ = 500
2.00 1.75 1.50
ERROR (dB)
2.0 1.5 1.0 0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
2.0 1.5 1.0 0.5 0 -0.5 -1.0 -1.5 -2.0 -50 -40 -30 -20 -10 0 10 PIN (dBm)
1.25
VOUT (V)
1.00 0.75 0.50 0.25 0 -60
1.00 0.75 0.50 0.25
Figure 11. VOUT and Log Conformance vs. Input Amplitude at 2.2 GHz, Multiple Devices, RTADJ = 8 k
05705-011
0 -60
Figure 14. VOUT and Log Conformance vs. Input Amplitude at 8.0 GHz, Multiple Devices, RTADJ =Open, Error Calculated from PIN = -34 dBm to PIN = -16 dBm
Rev. 0 | Page 8 of 20
05705-012
AD8319
j1 j0.5 j2
10000
j0.2
NOISE SPECTRAL DENSITY (nV/ Hz)
-60dBm 1000 RF OFF -20dBm 100 -40dBm -10dBm
0
0.2
0.5
1
2
100MHz -j0.2 900MHz 1900MHz 8000MHz -j0.5 -j2 -j1
05705-015
3600MHz START FREQUENCY = 0.05GHz STOP FREQUENCY = 10GHz 10000MHz
10 1k
10k
100k FREQUENCY (Hz)
1M
10M
5800MHz
Figure 15. Input Impedance vs. Frequency; No Termination Resistor on INHI (Impedance De-Embedded to Input Pins), Z0 = 50
: 1.53V @ : 1.53V
Figure 18. Noise Spectral Density of Output; CLPF = Open
10000
NOISE SPECTRAL DENSITY (nV/ Hz)
1000
100
1
05705-016
Ch1 500mV
M2.00s T 29.60%
A CH1
420V
10 1k
10k
100k FREQUENCY (Hz)
1M
10M
Figure 16. Power On/Off Response Time; VP = 3.0 V; Input AC-Coupling Caps = 10 pF; CLPF = Open
Figure 19. Noise Spectral Density of Output Buffer (from CLPF to VOUT); CLPF = 0.1 F
2.00 2.0 3.3V 1.75 1.50 1.25 3.0V 3.6V 1.5 1.0
CH1 RISE 9.949ns CH1 FALL 6.032ns
1.00 0.75 0.50 0.25
0 -0.5 -1.0 -1.5 -2.0 0 5 10
05705-017
Ch1 200mV
M20.0ns T 72.40%
A CH1
1.04V
PIN (dBm)
Figure 17. VOUT Pulse Response Time; Pulsed RF Input 0.1 GHz, -10 dBm; CLPF = Open; RLOAD = 150
Figure 20. Output Voltage Stability vs. Supply Voltage at 1.9 GHz When VPOS Varies by 10%
Rev. 0 | Page 9 of 20
05705-020
1
0 -60 -55 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5
ERROR (dB)
0.5
VOUT (V)
05705-019
05705-018
2200MHz
0dBm
AD8319 THEORY OF OPERATION
The AD8319 is a 5-stage demodulating logarithmic amplifier, specifically designed for use in RF measurement and power control applications at frequencies up to 10 GHz. A block diagram is shown in Figure 21. Sharing much of its design with the AD8318 logarithmic detector/controller, the AD8319 maintains tight intercept variability vs. temperature over a 40 dB range. Additional enhancements over the AD8318, such as reduced RF burst response time of 8 ns to 10 ns, 22 mA supply current, and board space requirements of only 2 mm x 3 mm add to the low cost and high performance benefits found in the AD8319.
VPSO TADJ
The logarithmic function is approximated in a piecewise fashion by five cascaded gain stages. (For a more comprehensive explanation of the logarithm approximation, please refer to the AD8307 data sheet, available at www.analog.com.) The cells have a nominal voltage gain of 9 dB each and a 3 dB bandwidth of 10.5 GHz. Using precision biasing, the gain is stabilized over temperature and supply variations. The overall dc gain is high due to the cascaded nature of the gain stages. An offset compensation loop is included to correct for offsets within the cascaded cells. At the output of each of the gain stages, a squarelaw detector cell is used to rectify the signal. The RF signal voltages are converted to a fluctuating differential current having an average value that increases with signal level. Along with the five gain stages and detector cells, an additional detector is included at the input of the AD8319, providing a 40 dB dynamic range in total. After the detector currents are summed and filtered, the following function is formed at the summing node: ID x log10(VIN/VINTERCEPT) where: ID is the internally set detector current. VIN is the input signal voltage. VINTERCEPT is the intercept voltage (that is, when VIN = VINTERCEPT, the output voltage would be 0 V, if it were capable of going to 0 V).
GAIN BIAS
SLOPE
V
I
VSET
I DET INHI INLO DET DET DET
V
VOUT
CLPF
COMM
Figure 21. Block Diagram
A fully differential design, using a proprietary, high speed SiGe process, extends high frequency performance. Input INHI receives the signal with a low frequency impedance of nominally 500 in parallel with 0.7 pF. The maximum input with 1 dB log-conformance error is typically 0 dBm (re: 50 ). The noise spectral density referred to the input is 1.15 nV/Hz, which is equivalent to a voltage of 118 V rms in a 10.5 GHz bandwidth or a noise power of -66 dBm (re: 50 ). This noise spectral density sets the lower limit of the dynamic range. However, the low end accuracy of the AD8319 is enhanced by specially shaping the demodulating transfer characteristic to partially compensate for errors due to internal noise. The common pin, COMM, provides a quality low impedance connection to the printed circuit board (PCB) ground. The package paddle, which is internally connected to the COMM pin, should also be grounded to the PCB to reduce thermal impedance from the die to the PCB.
Rev. 0 | Page 10 of 20
05705-021
AD8319 USING THE AD8319
BASIC CONNECTIONS
The AD8319 is specified for operation up to 10 GHz, as a result, low impedance supply pins with adequate isolation between functions are essential. A power supply voltage of between 3.0 V and 5.5 V should be applied to VPOS. Power supply decoupling capacitors of 100 pF and 0.1 F should be connected close to this power supply pin.
VS(2.7V-5.5V) C5 0.1F C4 100pF R2 0 SEE TEXT VOUT 47nF 8 INLO R1 52.3 INHI 1 7 VPOS 6 TADJ 5 VOUT R4 0 VSET 4
05705-022
combines with the relatively high input impedance to give an adequate broadband 50 match. The coupling time constant, 50 x CC/2, forms a high-pass corner with a 3 dB attenuation at fHP = 1/(2 x 50 x CC ), where C1 = C2 = CC. Using the typical value of 47 nF, this high-pass corner will be ~68 kHz. In high frequency applications, fHP should be as large as possible to minimize the coupling of unwanted low frequency signals. In low frequency applications, a simple RC network forming a low-pass filter should be added at the input for similar reasons. This should generally be placed at the generator side of the coupling capacitors, thereby lowering the required capacitance value for a given high-pass corner frequency.
C2
OUTPUT INTERFACE
The VOUT pin is driven by a PNP output stage. An internal 10 resistor is placed in series with the output and the VOUT pin. The rise time of the output is limited mainly by the slew on CLPF. The fall time is an RC-limited slew given by the load capacitance and the pull-down resistance at VOUT. There is an internal pull-down resistor of 1.6 k. A resistive load at VOUT is placed in parallel with the internal pull-down resistor to provide additional discharge current.
VPOS CLPF 10 + 0.8V - VOUT
AD8319
C1 47nF COMM 2 CLPF 3
SIGNAL INPUT
SEE TEXT
Figure 22. Basic Connections
The paddle of the LFCSP package is internally connected to COMM. For optimum thermal and electrical performance, the paddle should be soldered to a low impedance ground plane.
INPUT SIGNAL COUPLING
The RF input (INHI) is single-ended and must be ac-coupled. INLO (input common) should be ac-coupled to ground. Suggested coupling capacitors are 47 nF ceramic 0402-style capacitors for input frequencies of 1 MHz to 10 GHz. The coupling capacitors should be mounted close to the INHI and INLO pins. The coupling capacitor values can be increased to lower the input stage's high-pass cutoff frequency. The highpass corner is set by the input coupling capacitors and the internal 10 pF high-pass capacitor. The dc voltage on INHI and INLO is about one diode voltage drop below VPOS.
VPOS 5pF 5pF FIRST GAIN STAGE 2k INLO Gm STAGE OFFSET COMP
05705-023
1200 400
05705-024
COMM
Figure 24. Output Interface
To reduce the fall time, VOUT should be loaded with a resistive load of <1.6 k. For example, with an external load of 150 the AD8319 fall time is <7 ns.
SETPOINT INTERFACE
The VSET input drives the high impedance (20 k) input of an internal op amp. The VSET voltage appears across the internal 1.5 k resistor to generate ISET. When a portion of VOUT is applied to VSET, the feedback loop forces -ID x log10(VIN/VINTERCEPT) = ISET. If VSET = VOUT/2x, then ISET = VOUT/(2x x 1.5 k). The result is VOUT = (-ID x 1.5 k x 2x) x log10(VIN/VINTERCEPT)
CURRENT
18.7k INHI
18.7k
A = 9dB
Figure 23. Input Interface
While the input can be reactively matched, in general this is not necessary. An external 52.3 shunt resistor (connected on the signal side of the input coupling capacitors, as shown in Figure 22)
Rev. 0 | Page 11 of 20
AD8319
20k VSET VSET ISET
MEASUREMENT MODE
When the VOUT voltage or a portion of the VOUT voltage is fed back to the VSET pin, the device operates in measurement mode. As seen in Figure 27, the AD8319 has an offset voltage, a negative slope, and a VOUT measurement intercept at the high end of its input signal range.
2.00 1.75 1.50 1.25
VOUT (V)
20k 1.5k COMM COMM
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Figure 25. VSET Interface
2.0 VOUT 25C ERROR 25C 1.5 1.0 0.5 0 -0.5 -1.0 -1.5
05705-027
The slope is given by -ID x 2x x 1.5 k = -22 mV/dB x x. For example, if a resistor divider to ground is used to generate a VSET voltage of VOUT/2, then x = 2. The slope is set to -880 V/decade or -44 mV/dB.
TEMPERATURE COMPENSATION OF OUTPUT VOLTAGE
The primary component of the variation in VOUT vs. temperature, as the input signal amplitude is held constant, is the drift of the intercept. This drift is also a weak function of the input signal frequency, so provision is made for optimization of internal temperature compensation at a given frequency by providing Pin TADJ.
VINTERNAL
1.00 0.75 0.50 0.25 RANGE FOR CALCULATION OF SLOPE AND INTERCEPT
0 -60 -55 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 PIN (dBm)
0
5
10 15 INTERCEPT
AD8319
ICOMP
Figure 27. Typical Output Voltage vs. Input Signal
TADJ RTADJ
The output voltage vs. input signal voltage of the AD8319 is linear-in-dB over a multidecade range. The equation for this function is of the form VOUT = X x VSLOPE/DEC x log10(VIN/VINTERCEPT) = (1) (2)
05705-026
1.5k COMM COMM
X x VSLOPE/dB x 20 x log10(VIN/VINTERCEPT) where: X is the feedback factor in VSET = VOUT/X. VSLOPE/DEC is nominally -440 mV/decade or -22 mV/dB. VINTERCEPT is the x-axis intercept of the linear-in-dB portion of the VOUT vs. VIN curve (Figure 27). VINTERCEPT is +2 dBV for a sinusoidal input signal. An offset voltage, VOFFSET, of 0.35 V is internally added to the detector signal, so that the minimum value for VOUT is X x VOFFSET, so for X = 1, minimum VOUT is 0.35 V.
Figure 26. TADJ Interface
The Resistor RTADJ is connected between this pin and ground. The value of this resistor partially determines the magnitude of an analog correction coefficient, which is used to reduce intercept drift. The relationship between output temperature drift and frequency is not linear and cannot be easily modeled. As a result, experimentation is required to choose the correct TADJ resistor. Table 4 shows the recommended values for some commonly used frequencies. Table 4. Recommended RTADJ Resistor Values
Frequency 50 MHz 100 MHz 900 MHz 1.8 GHz 1.9 GHz 2.2 GHz 3.6 GHz 5.3 GHZ 5.8 GHz 8 GHz Recommended RTADJ 18 k 18 k 18 k 8 k 8 k 8 k 8 k 500 500 Open
The slope is very stable vs. process and temperature variation. When base-10 logarithms are used, VSLOPE/DECADE represents the volts/decade. A decade corresponds to 20 dB; VSLOPE/DECADE/20 = VSLOPE/dB represents the slope in volts/dB. As noted in the Equation 1 and Equation 2, the VOUT voltage has a negative slope. This is also the correct slope polarity to control the gain of many power amplifiers in a negative feedback configuration. Because both the slope and intercept vary slightly with frequency, it is recommended to refer to the Specifications section for application-specific values for slope and intercept.
Rev. 0 | Page 12 of 20
AD8319
Although demodulating log amps respond to input signal voltage, not input signal power, it is customary to discuss the amplitude of high frequency signals in terms of power. In this case, the characteristic impedance of the system, Z0, must be known to convert voltages to their corresponding power levels. The following equations are used to perform this conversion. P(dBm) = 10 x log10(Vrms2/(Z0 x 1 mW)) P(dBV) = 20 x log10(Vrms/1 Vrms) P(dBm) = P(dBV) - 10 x log10(Z0 x 1 mW/1 Vrms2) (3) (4) (5)
CONTROLLER MODE
The AD8319 provides a controller mode feature at the VOUT pin. Using VSET for the setpoint voltage, it is possible for the AD8319 to control subsystems, such as power amplifiers (PAs), variable gain amplifiers (VGAs), or variable voltage attenuators (VVAs) that have output power that increases monotonically with respect to their gain control signal. To operate in controller mode, the link between VSET and VOUT is broken. A setpoint voltage is applied to the VSET input; VOUT is connected to the gain control terminal of the variable gain amplifier (VGA) and the detector's RF input is connected to the output of the VGA (usually using a directional coupler and some additional attenuation). Based on the defined relationship between VOUT and the RF input signal when the device is in measurement mode, the AD8319 adjusts the voltage on VOUT (VOUT is now an error amplifier output) until the level at the RF input corresponds to the applied VSET. When the AD8319 operates in controller mode, there is no defined relationship between VSET and VOUT voltage; VOUT settles to a value that results in the correct input signal level appearing at INHI/INLO. For this output power control loop to be stable, a groundreferenced capacitor must be connected to the CLPF pin. This capacitor, CFLT, integrates the error signal (in the form of a current) to set the loop bandwidth and ensure loop stability. Further details on control loop dynamics can be found in the AD8315 data sheet.
For example, PINTERCEPT for a sinusoidal input signal expressed in terms of dBm (decibels referred to 1 mW), in a 50 system is PINTERCEPT(dBm) = PINTERCEPT(dBV) - 10 x log10(Z0 x 1 mW/1 Vrms2) = +2 dBV - 10 x log10(50x10-3) = +15 dBm For a square wave input signal in a 200 system PINTERCEPT = -1 dBV - 10 x log10[(200 x 1 mW/1Vrms2)] = +6 dBm Further information on the intercept variation dependence upon waveform can be found in the AD8313 and AD8307 data sheets. (6)
SETTING THE OUTPUT SLOPE IN MEASUREMENT MODE
To operate in measurement mode, VOUT must be connected to VSET. Connecting VOUT directly to VSET yields the nominal logarithmic slope of -22 mV/dB. The output swing corresponding to the specified input range is then 0.35 V to 1.5 V. The slope and output swing can be increased by placing a resistor divider between VOUT and VSET (that is, one resistor from VOUT to VSET and one resistor from VSET to ground). The input impedance of VSET is 40 k. Slope-setting resistors should be kept below 20 k to prevent this input impedance from affecting the resulting slope. If two equal resistors are used (for example, 10 k/10 k), the slope doubles to -44 mV/dB.
AD8319
VOUT 10k VSET
05705-028
VGA/VVA DIRECTIONAL COUPLER ATTENUATOR 47nF INHI 52.3 GAIN CONTROL VOLTAGE VOUT
RFIN
AD8319
INLO 47nF VSET
DAC
CLPF CFLT
05705-029
-44mV/dB
Figure 29. AD8319 Controller Mode
10k
Figure 28. Increasing the Slope
Decreasing VSET, which corresponds to demanding a higher signal from the VGA, increases VOUT. The gain control voltage of the VGA must have a positive sense. A positive control voltage to the VGA increases the gain of the device.
Rev. 0 | Page 13 of 20
AD8319
The basic connections for operating the AD8319 in an automatic gain control (AGC) loop with the ADL5330 are shown in Figure 30. The ADL5330 is a 10 MHz to 3 GHz variable gain amplifier. It offers a large gain control range of 60 dB with 0.5 dB gain stability. This configuration is similar to Figure 29. The gain of the ADL5330 is controlled by the output pin of the AD8319. This voltage, VOUT, has a range of 0 V to near VPOS. To avoid overdrive recovery issues, the AD8319 output voltage can be scaled down using a resistive divider to interface with the 0 V to 1.4 V gain control range of the ADL5330. A coupler/attenuation of 21 dB is used to match the desired maximum output power from the VGA to the top end of the linear operating range of the AD8319 (approximately -5 dBm at 900 MHz).
+5V RF INPUT SIGNAL VPOS 100pF INHI OPHI COMM 120nH 100pF +5V RF OUTPUT SIGNAL
30 20 10 0 -10 -20 -30 -40
4 3 2
OUTPUT POWER (dBm)
0 -1 -2 -3
SETPOINT VOLTAGE (V)
Figure 31. ADL5330 Output Power vs. AD8319 Setpoint Voltage, PIN = -1.5 dBm
120nH
ADL5330
INLO 100pF GAIN OPLO
100pF
DIRECTIONAL COUPLER
4.12k 10k SETPOINT VOLTAGE DAC VOUT VSET
+5V
ATTENUATOR
VPOS INHI
47nF 52.3
AD8319
LOG AMP
1nF
CLPF TADJ
INLO COMM
47nF
For the AGC loop to remain in equilibrium, the AD8319 must track the envelope of the ADL5330's output signal and provide the necessary voltage levels to the ADL5330's gain control input. Figure 32 shows an oscilloscope screenshot of the AGC loop depicted in Figure 30. A 100 MHz sine wave with 50% AM modulation is applied to the ADL5330. The output signal from the VGA is a constant envelope sine wave with amplitude corresponding to a setpoint voltage at the AD8319 of 1.3 V. Also shown is the gain control response of the AD8319 to the changing input envelope.
05705-030
18k
AM MODULATED INPUT
Figure 30. AD8319 Operating in Controller Mode to Provide Automatic Gain Control Functionality in Combination with the ADL5330
1
Figure 31 shows the transfer function of the output power vs. the VSET voltage over temperature for a 900 MHz sine wave with an input power of -1.5 dBm. Note that the power control of the AD8319 has a negative sense. Decreasing VSET, which corresponds to demanding a higher signal from the ADL5330, increases gain.
3
AD8319 OUTPUT
The AGC loop is capable of controlling signals of ~40 dB. This range limitation is due to the dynamic range of the AD8319. Using a wider dynamic range detector such as the AD8317, AD8318, or AD8362 will allow for the full 60dB range of the ADL5330 to be utilized. The performance over temperature is most accurate over the highest power range, where it is generally most critical. Across the top 40 dB range of output power, the linear conformance error is well within 0.5 dB over temperature.
2 ADL5330 OUTPUT CH1 200mV Ch2 200mV Ch3 100mV
M2.00ms A Ch2 T 0.00000 s
1.03V
Figure 32. Oscilloscope Screenshot Showing an AM Modulated Input Signal and the Response from the AD8319
Figure 33 shows the response of the AGC RF output to a pulse on VSET. As VSET decreases from 1.5 V to 0.4 V, the AGC loop responds with an RF burst. In this configuration the input signal to the ADL5330 is a 1 GHz sine wave at a power level of -15 dBm.
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05705-032
05705-031
-4 -50 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.6
ERROR (dB)
1
AD8319
T
CFLT is selected using the following equation:
AD8319 VSET PULSE
CFLT =
1
( x 1.5 k x Video Bandwidth )
1
- 3.5 pF (10)
ADL5330 OUTPUT
3
The video bandwidth should typically be set to a frequency equal to about one-tenth the minimum input frequency. This ensures that the output ripple of the demodulated log output, which is at twice the input frequency, is well filtered. In many log amp applications, it may be necessary to lower the corner frequency of the postdemodulation filtering to achieve low output ripple while maintaining a rapid response time to changes in signal level. An example of a 4-pole active filter is shown in the AD8307 data sheet.
Ch1 2.00V Ch3 50mV
M10.0s A Ch1 T 179.800s
2.60V
Figure 33. Oscilloscope Screenshot Showing the Response Time of the AGC Loop
05705-033
OPERATION BEYOND 8 GHZ
The AD8319 is specified for operation up to 8 GHz, but it provides useful measurement accuracy over a reduced dynamic range of up to 10 GHz. Figure 35 shows the performance of the AD8319 over temperature at 10 GHz when the device is configured as shown in Figure 22. Dynamic range is reduced at this frequency, but the AD8319 does provide 30 dB of measurement range within 3 dB of linearity error.
2.0 1.8 1.6 1.4
VOUT (V)
Response time and the amount of signal integration are controlled by CFLT. This functionality is analogous to the feedback capacitor around an integrating amplifier. While it is possible to use large capacitors for CFLT, in most applications values under 1 nF provide sufficient filtering. Calibration in controller mode is similar to the method used in measurement mode. A simple two-point calibration can be done by applying two known VSET voltages or DAC codes and measuring the output power from the VGA. Slope and intercept can then be calculated with the following equations: Slope = (VSET1 - VSET2)/(POUT1 - POUT2) Intercept = POUT1 - VSET1/Slope VSETX = Slope x (POUTX - Intercept) (7) (8) (9)
5 4 3 2 1 0 -1 -2 -3 -4
05705-038
1.2 1.0 0.8 0.6 0.4 0.2
More information on the use of the ADL5330 in AGC applications can be found in the ADL5330 data sheet.
OUTPUT FILTERING
For applications in which maximum video bandwidth and, consequently, fast rise time are desired, it is essential that the CLPF pin be left unconnected and free of any stray capacitance. The nominal output video bandwidth of 50 MHz can be reduced by connecting a ground-referenced capacitor (CFLT) to the CLPF pin, as shown in Figure 34. This is generally done to reduce output ripple (at twice the input frequency for a symmetric input waveform such as sinusoidal signals).
AD8319
ILOG +4 1.5k 3.5pF VOUT
0 -40
-5 -35 -30 -25 -20 -15 PIN (dBm) -10 -5 0 5
Figure 35. VOUT and Log Conformance vs. Input Amplitude at 10.0 GHz, Multiple Devices, RTADJ = Open, CLPF = 1000 pF
Implementing an impedance match for frequencies beyond 8 GHz can improve the sensitivity of the AD8319 and measurement range. Operation beyond 10 GHz is possible, but part to part variation, most notably in the intercept, becomes significant.
CLPF CFLT
05705-037
Figure 34. Lowering the Postdemodulation Bandwidth
Rev. 0 | Page 15 of 20
ERROR (dB)
AD8319 EVALUATION BOARD
Table 5. Evaluation Board (Rev. A) Configuration Options
Component VPOS, GND R1, C1, C2 Function Supply and Ground Connections. Input Interface. The 52.3 resistor in position R1 combines with the AD8319's internal input impedance to give a broadband input impedance of about 50 . Capacitor C1 and Capacitor C2 are dc-blocking capacitors. A reactive impedance match can be implemented by replacing R1 with an inductor and C1 and C2 with appropriately valued capacitors. Temperature Compensation Interface. The internal temperature compensation network is optimized for input signals up to 3.6 GHz when R7 is 10 k. This circuit can be adjusted to optimize performance for other input frequencies by changing the value of the resistor in position R7. See Table 4 for specific TADJ resistor values. Output Interface--Measurement Mode. In measurement mode, a portion of the output voltage is fed back to Pin VSET via R2. The magnitude of the slope of the VOUT output voltage response can be increased by reducing the portion of VOUT that is fed back to VSET. R6 can be used as a back-terminating resistor or as part of a single-pole low-pass filter. Default Conditions Not applicable R1 = 52.3 (Size 0402) C1 = 47 nF (Size 0402) C2 = 47 nF (Size 0402) R5 = 200 (Size 0402) R7 = open (Size 0402) R2 = 0 (Size 0402) R3 = open (Size 0402) R4 = open (Size 0402) R6 = 1 k (Size 0402) RL = CL = open (Size 0402) R2 = open (Size 0402) R3 = open (Size 0402)
R5, R7
R2, R3, R4, R6, RL, CL
R2, R3
C4, C5,
Output Interface--Controller Mode. In this mode, R2 must be open. In controller mode, the AD8319 can control the gain of an external component. A setpoint voltage is applied to Pin VSET, the value of which corresponds to the desired RF input signal level applied to the AD8319 RF input. A sample of the RF output signal from this variable-gain component is selected, typically via a directional coupler, and applied to AD8319 RF input. The voltage at Pin VOUT is applied to the gain control of the variable gain element. A control voltage is applied to Pin VSET. The magnitude of the control voltage can optionally be attenuated via the voltage divider comprising R2 and R3, or a capacitor can be installed in position R3 to form a low-pass filter along with R2. Power Supply Decoupling. The nominal supply decoupling consists of a 100 pF filter capacitor placed physically close to the AD8319 and a 0.1 F capacitor placed nearer to the power supply input pin. Filter Capacitor. The low-pass corner frequency of the circuit that drives Pin VOUT can be lowered by placing a capacitor between CLPF and ground. Increasing this capacitor increases the overall rise/fall time of the AD8319 for pulsed input signals. See the Output Filtering section for more details.
C3
C5 = 100 pF (Size 0402) C4 = 0.1 F (Size 0603) C3 = 8.2 pF (Size 0402)
Rev. 0 | Page 16 of 20
AD8319
VPOS C4 0.1F C5 100pF R7 OPEN R5 200 VOUT_ALT R4 OPEN R6 1k 8 INLO R1 52.3 INHI 1 7 VPOS 6 TADJ 5 VOUT R2 0 VSET 4 VSET
0705-034
TADJ GND
C1 47nF
CL OPEN
RL OPEN
VOUT
AD8319
C2 47nF COMM 2 CLPF 3
RFIN
C3 8.2pF
R3 OPEN
Figure 36. Evaluation Board Schematic (Rev. A)
Figure 37. Component Side Layout
05705-035
Figure 38. Component Side Silkscreen
Rev. 0 | Page 17 of 20
05705-036
AD8319 OUTLINE DIMENSIONS
3.25 3.00 2.75 2.25 2.00 1.75 0.60 0.45 0.30 1.89 1.74 1.59
5 BOTTOM VIEW 8 * EXPOSED PAD 4 1
0.55 0.40 0.30
1.95 1.75 1.55
TOP VIEW
0.15 0.10 0.05 0.25 0.20 0.15
PIN 1 INDICATOR
2.95 2.75 2.55 12 MAX 0.80 MAX 0.65 TYP
0.50 BSC
1.00 0.85 0.80
0.05 MAX 0.02 NOM 0.30 0.23 0.18 0.20 REF
SEATING PLANE
Figure 39: 8-Lead Lead Frame Chip Scale Package [LFCSP_VD] 2 mm x 3 mm Body, Very Thin, Dual Lead (CP-8-1) Dimensions shown in millimeters
ORDERING GUIDE
Model AD8319ACPZ-R71 AD8319ACPZ-R21 AD8319ACPZ-WP1, 2 AD8319-EVAL
1 2
Temperature Range -40C to +85C -40C to +85C -40C to +85C
Package Description 8-Lead LFCSP_VD 8-Lead LFCSP_VD 8-Lead LFCSP_VD Evaluation Board
Package Option CP-8-1 CP-8-1 CP-8-1
Branding Q2 Q2 Q2 Q2
Z = Pb-free part. WP = waffle pack.
Rev. 0 | Page 18 of 20
AD8319 NOTES
Rev. 0 | Page 19 of 20
AD8319 NOTES
(c)2005 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D05705-0-10/05(0)
Rev. 0 | Page 20 of 20

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